Damping or resonant peaks in an electric motor which is operated using a converter with an intermediate voltage circuit, by means of a transformer-coupled damping resistance, and a corresponding electric motor

ABSTRACT

In a converter system having an intermediate voltage circuit which operates with a supply network-side input inductor in the step-converter mode or has other input-side inductances, there is a risk of natural system oscillations being formed via discharge capacitances in conjunction with motors. If the motor now has an amplitude/frequency response with a pronounced resonant frequency in the region of such natural system oscillations, then there is a risk of higher voltages occurring at the motor star point (S) than in the motor phases (I, V, W). This is prevented by the present invention by introducing an impedance (Z), in particular at the input to the motor, in order to damp capacitive discharge currents to ground potential, which are caused by system oscillations (f sys ) (excited asymmetrically with respect to ground in the motor phases (U, V, W)) of the converter system (L K , UR, LT, M) in the winding sections.

FIELD OF THE INVENTION

[0001] The invention relates to a method for damping resonant peaks at amotor star point in an electric motor which is operated using anintermediate voltage circuit converter with an input-side inductance, inparticular a mains system input indicator (supply network-side inputinductor,) and which, owing to characteristics of its winding sections,has a frequency response with at least one resonant frequency withrespect to ground potential, and to a corresponding electric motor inwhich resonant peaks are damped in such a manner.

BACKGROUND OF THE INVENTION

[0002] In present-day converter systems having an intermediate voltagecircuit, in particular in multi-shaft converter systems of this type,system oscillations can be formed which are virtually undamped. This isespecially true in converters having an intermediate voltage circuit anda regulated supply in the form of a regulated supply network-sideconverter, which is also referred to as an input converter.

[0003] Converters of this type are used for operating electricalmachines with a variable supply frequency. Such an intermediate circuitfrequency converter allows an electric motor, for example a three-phasemachine such as a synchronous machine, to be operated not only in such amanner that it is linked directly to the supply network and hence has afixed rotation speed, but also such that the fixed supply network can bereplaced by an electronically produced, variable-frequency andvariable-amplitude supply for powering the electrical machine.

[0004] The two supply systems, (i.e. the supply network whose amplitudeand frequency are fixed, and the supply system which supplies theelectrical machine with a variable amplitude and frequency), are coupledvia a DC voltage storage device or a DC current storage device in theform of what is referred to as an intermediate circuit. In this case,such intermediate circuit converters essentially have three centralassemblies:

[0005] a supply network-side input converter which can be designed to beunregulated (for example diode bridges) or to be regulated, in whichcase energy can be fed back into the supply network only by using aregulated input converter;

[0006] an energy storage device in the intermediate circuit in the formof a capacitor in the case of an intermediate voltage circuit and aninductor in the case of an intermediate current circuit; and

[0007] an output-side converter or inverter for supplying the machine,which generally uses a three-phase bridge circuit having six activecurrent devices which can be turned off, for example IGBT transistors,to convert the DC voltage in an intermediate voltage circuit into athree-phase voltage system.

[0008] Such a converter system having an intermediate voltage circuitwhich, inter alia, owing to its very wide frequency and amplitudecontrol range, is preferably used for main drives and servo drives inmachine tools, robots and production machines, is shown in theillustration in FIG. 1.

[0009] The converter UR is connected to a three-phase supply network Nvia filter F and an energy-storage inductor whose inductance is L_(K).The converter UR has the described converted E, an intermediate voltagecircuit with an energy-storage capacitance C_(ZK), and an outputinverter W. FIG. 1 shows a regulated converter E which is operated suchthat it is controlled by switching components (for example a three-phasebridge circuit composed of IGBT transistors), as a result of which thearrangement experiences excitation A1. The inverter W is likewisecontrolled via further switching components, for example, by means of athree-phase bridge circuit having six IGBT transistors. As a result ofthose switching operations the inverter W experiences excitation A2 ofthe system. The capacitor C_(ZK) in the intermediate voltage circuit isconnected between the positive intermediate circuit rail P600 and thenegative intermediate circuit rail M600. The inverter is connected onthe output side via a line LT, having a protective-ground conductor PEand a shield SM, to a motor M in the form of a three-phase machine.

[0010] The fixed-frequency three-phase supply network N now supplies theintermediate circuit capacitor C_(ZK) via the input converter E and viathe filter F and the energy-storage inductor L_(K) by means of theregulated supply, with the input converter E (for example apulse-controlled converter) operating together with the energy-storageinductor L_(K) as a step-up converter. Once current flows into theenergy-storage inductor L_(K), it is connected to the intermediatecircuit and drives the current into the capacitor C_(ZK). Theintermediate circuit voltage may therefore be greater than the peakvalue of the supply network voltage.

[0011] This combination effectively represents a DC voltage source. Theinverter W uses this DC voltage in the described manner to form athree-phase voltage system in which case, in contrast to the sinusoidalvoltage of a three-phase generator, the output voltage does not have theprofile of an ideal sinusoidal oscillation, but also has harmonics inaddition to the fundamental, since it is produced electronically via abridge circuit.

[0012] However, in addition to the described elements in such anarrangement, it is also necessary to consider parasitic capacitanceswhich assist the formation of system oscillations in such a convertersystem. Thus, in addition to the filter F with the discharge capacitanceC_(F), the input converter E, the inverter W and the motor M also havedischarge capacitances C_(E), C_(W) and C_(M) to ground. Furthermore,there is also a capacitance C_(PE) in the line LT to theprotective-ground conductor PE, and a capacitance C_(SM) in the line LTto the grounded shield SM.

[0013] It has now been found that these system oscillations are excitedto a particularly pronounced extent in the converter E. Depending on thecontrol method chosen for the supply, two or three phases of the supplynetwork N are short-circuited, in order to pass current to theenergy-storage inductor L_(K). If all three phases U, V, W areshort-circuited, then either the positive intermediate circuit rail P600or the negative intermediate circuit rail M600 is hard-connected to thestar point of the supply network (generally close to ground potentialdepending on the zero phase-sequence system component). If two phases ofthe supply network N are short-circuited, then the relevant intermediatecircuit rails P600 and M600 are hard-connected to an inductive voltagedivider between the two supply network phases.

[0014] Depending on the situation relating to the supply networkvoltages, this voltage is in the vicinity of ground potential(approximately 50-60 V). Since the intermediate circuit capacitanceC_(ZK) is generally large (continuous voltage profile), the otherintermediate circuit rail is 600 V lower or higher and can thus alsobreak down the remaining phase of the supply network. In both cases, theintermediate circuit is particularly severely deflected from its“natural” balanced steady-state position (±300 V with respect toground), which represents a particularly severe excitation for systemoscillation.

[0015] With respect to the production of undesirable systemoscillations, the frequency band below 50 to 100 kHz area, which isrelevant for the application, allows a resonant frequency to becalculated based on concentrated elements. In this case, the dischargecapacitances C_(F) to ground in the filter F are generally so large thatthey do not have a frequency-governing effect. In this case, it can beassumed that dominant excitation to oscillations takes place upstream ofthe described capacitances, and that the filter discharge capacitanceC_(F) can be ignored.

[0016] The resonant frequency f_(res)(sys) of this system, which isreferred to as f_(sys) in the following text, is thus given by:

[0017] where $\begin{matrix}{{f_{sys} = \frac{1}{2\pi \sqrt{L_{\sum} \cdot C_{\sum}}}}{where}} & (1) \\{L_{\sum} = {L_{K} + L_{F}}} & (2)\end{matrix}$

[0018] where L_(K) represents the dominant component and L_(F) theunbalanced inductive elements acting on the converter side in the filter(for example current-compensated inductors); and

C _(Σ) =C _(E) +C _(W) +C _(PE) +C _(SM) +C _(M)  (3)

[0019] This relationship is shown in FIG. 2. In this case, L_(Σ) andC_(Σ) form a passive circuit, which is excited by excitation A andstarts to oscillate at its natural resonant frequency f_(sys).

[0020] Accordingly, in addition to the shifts with an amplitude of 600V, for example, that occur during operation, an additional, undesirableoscillation with an amplitude of up to several hundred volts is alsomodulated onto the voltages of the intermediate circuit rails P600 andM600.

[0021] In electric motors M in general, but in particular if they aredesigned using field coil technology (for example torque motors), afrequency response with pronounced resonant peaks with respect to groundpotential can occur if such motors are excited in the common mode withrespect to ground at all the motor terminals, for example due to theundesirable system oscillations described above.

[0022] These resonant points can be explained by an unbalancedequivalent circuit comprising a lattice network circuit K of parasiticelements (inductances L and discharge capacitances C) in the motorwinding, as in FIG. 3 which shows the winding section of one phase U ofa three-phase motor M with the three phases U, V, W whose windingsections are electrically connected to one another at the motor starpoint S. The input voltages of the three-phase current generated by theinverter W are applied to the outer terminals, which are opposite thestar point S, of the respective winding sections.

[0023] This applies in particular to motors using field coil technology,in which individual four-pole networks in the lattice network K arepossible by virtue of the construction, and essentially correspond to asingle field coil. In field coil technology, the magnetic cores, whichare composed of magnetic steel laminates, have teeth which act as polecores, which are placed onto the prefabricated coils and are connectedas appropriate. The individual inductances L are, as can be seen in FIG.3, electrically connected in series, with each field coil beingcapacitively coupled to the pole core (magnetic steel laminate) on whichthe coil is fit. These respective capacitances are represented asdischarge capacitances C to ground, which are formed by the magneticcore.

[0024] However, the described phenomenon can also be explained in thecase of motors with a different construction (for example using what isreferred to as wild winding) by a model of a lattice network K, by thismodel representing an equivalent circuit with identical four-polenetworks in the form of LC tuned circuits, whose elements simulate thefrequency response. The peak in this case occurs in the region of thestar point S, which is normally not deliberately subjected to voltageloads. If the system oscillation of the converter system occurs in thevicinity of a natural motor frequency, then the insulation system toground can be overloaded, in particular at the star point S, leading topremature failure of the motor M, since, due to resonance, considerablyhigher voltages can occur at the motor star point than at the motorterminals.

[0025] This is true for all voltage levels (low-voltage, medium-voltageand high-voltage systems), but particularly when the step-up converterprinciple is used (with the energy-storage inductor L_(K)) on theconverter side UR and a frequency response with pronounced resonantpeaks with respect to ground potential occurs on the other side in themotor M as is the case in motors with a particularly low natural motorfrequency because the natural damping in the motor resulting from eddycurrent losses and hysteresis losses etc. is particularly low.

[0026] Similar problems arise repeatedly in the field of electricalmachines when transient overvoltages occur. The overvoltages are thuslimited in order to avoid flashovers. For example, according to Germanpatent document DE-A-38 26 282, a voltage-dependent metal-oxide resistoris connected in parallel with a coil in order to limit overvoltages. InGerman patent document DE-B-28 34 378, winding sections areshort-circuited in order to damp quadrature-axis field. In a similarway, according to German patent document DE-A-24 33 618, transientovervoltages in a synchronous machine are damped by means ofquadrature-axis field damper bars.

[0027] Furthermore, European patent document EP-A-0 117 764 describeshow overvoltages which occur due to resonance phenomena can besuppressed by ferroelectric insulators between the coil windings.Finally, European patent document EP-B-0 681 361 addresses the problemof higher-order harmonic oscillations, which can occur in converters andrectifiers using power thyristors. The damper winding is in consequenceconnected to capacitors in order to form tuned circuits. The tunedcircuits have a resonant frequency which is six times as high as thefundamental frequency of the synchronous machine. Higher-order harmonicoscillations on a fimdamental can thus be absorbed. Nevertheless, theproblem of possible resonant peaks at the star point S of a motor Mstill remains.

SUMMARY OF THE INVENTION

[0028] It is an object of the present invention to avoid resonant peaksexcited by such system oscillations in an electric motor operated usingsuch a converter system. This object is achieved by a method for dampingresonant peaks at a motor star point in an electric motor which isoperated using an intermediate voltage circuit converter with aninput-side inductance, in particular a supply network-side inputinductor, and which, owing to characteristics of its winding sections,has a frequency response with at least one resonant frequency withrespect to ground potential, in that an impedance for damping capacitivedischarge currents to ground potential, which are caused in the windingsections, is introduced into all the motor phases leading to a motorstar point. For this purpose, this impedance is designed for capacitivedischarge currents which are caused by system oscillations (which areexcited asymmetrically with respect to ground in the motor phases) ofthe converter system.

[0029] It has also been found to be preferred if all the motor phasesleading to a motor star point are routed through a lossy magneticcoupling core. In this case, the losses required for damping canadvantageously be produced by the characteristics of the magneticmaterial itself. Alternatively, this can be achieved if the couplingcore, for example a magnetic core, has a winding which isshort-circuited via an impedance. In this case, it has been found to beadvantageous, particularly with regard to possible retrofitting ofmotors, for all the motor phases leading to a motor star point to berouted through the lossy magnetic core at the input to the motor. If theimpedance is a non-reactive resistance, then this results in aparticularly simple and cost-effective implementation.

[0030] On the basis of the knowledge that each winding section of themotor forms an LC lattice network, the resistance of the non-reactiveresistor is preferably determined by:$R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}$

[0031] where L is the inductance and C the discharge capacitance of onelattice network element in the LC lattice network structure.

[0032] The total inductance of the coupling circuit formed with themagnetic core is preferably given by:

[0033] where,${L_{H1} \geq \frac{R}{2\pi \quad f_{0}}},{where},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

[0034] with f_(res) being a pronounced resonant frequency in theamplitude/frequency response of the motor.

[0035] In this case, the method according to the present invention canalso be used for an electric motor having a number of motor star points,in particular for a linear motor or a torque motor, by carrying out themethod for each motor star point, and with a suitable impedance in eachcase being coupled into all the motor phases leading to a motor starpoint. In this case as well, the impedance can be transformed in at themotor input, in which case it is then sensible to route all the motorphases to all the motor star points through a single coupling core,which must be dimensioned as appropriate. A torque motor is a machinewhich is designed to produce high torques, generally at low rotationspeeds, for example in the form of a brushless synchronous motor with alarge number of poles and permanent-magnet excitation.

[0036] If the converter is operated together with a supply network inputinductor in order to provide a supply based on the step-up converterprinciple, then the invention results in significant advantages withregard to undamped system oscillations.

[0037] Furthermore, the object of the present invention is also achievedby an electric motor using an intermediate voltage circuit converterhaving an input-side inductance, in particular a supply network inputinductor, having a frequency response, which is governed bycharacteristics of its winding sections with at least one resonantfrequency with respect to ground potential, and in which all the motorphases leading to a motor star point are routed through a lossy magneticcoupling core. This can be easily achieved by using the characteristicsof the magnetic material itself to produce the losses required fordamping, or by the magnetic core having a winding which isshort-circuited via an impedance. It preferred for the magnetic core tobe arranged at the input to the motor. Since this impedance is designedfor damping capacitive discharge currents with respect to groundpotential which are caused by system oscillations (excitedasymmetrically with respect to ground in the motor phases) of theconverter system in the winding sections, system oscillations of theconverter system can be suppressed particularly effectively, togetherwith resonant peaks produced by them in the motor, on the basis of apassive circuit, particularly in conjunction with a supply network-sideinput inductor. It is particularly preferred for the impedance to be anon-reactive resistance. The value of this non-reactive resistance andof the total inductance is preferably governed by the same rules as forthe method according to the invention, where$R_{a} \geq {{\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}\quad {and}\quad L_{H1}} \geq {\frac{R}{2\pi \quad f_{0}}.}$

[0038] The success of the invention can in this case be improved furtherby the coupling core being constructed such that it does not entersaturation at any operating point of the motor. The solution proposed bythe present invention has been found to be particularly advantageous formotors with winding sections using field core technology, which eachform a lattice network structure composed of inductances L and dischargecapacitances C, with the impedance being used for transformer damping ofthese lattice network structures. This is achieved particularly well ifthe impedance is designed such that it damps common-mode currents, whichare excited asymmetrically with respect to ground in the motor phases,of the converter system in the lattice network structure.

[0039] However, the principle of the invention can also be applied toany other desired forms of electric motors, particularly also thoseusing what is referred to as wild winding technology, in particularlow-voltage motors. This has been found to be particularly advantageousfor such drives whose geometric dimensions are large and in which largeslot areas result in large discharge capacitances, which lead toparticularly low resonant frequencies f_(res). This is because the riskof resonant peaks at the motor star point is low provided suchpronounced resonant points of the motor are well above any possibleconverter system oscillations. However, the situation changes, thecloser such resonant frequencies in the frequency response of a motorwith respect to ground potential are in the region of such convertersystem oscillations. This relates primarily to the physical size of themotor. The size of a motor governs the slot area which itself affectsthe capacitance C_(M) of the motor with respect to ground potential,since this discharge capacitance increases with the size of the slotarea. As the discharge capacitance C_(M) of the motor increases, thepronounced resonant frequency f_(res) in the amplitude/frequencyresponse of the motor with respect to ground potential in turn falls andthus comes closer to the region of undesirable natural systemfrequencies f_(sys) of the converter system. This means that, as thegeometric dimensions of the motor, for example the physical length orthe diameter, increase, pronounced resonant frequencies come closer tothis critical region, and the problem of resonant peaks becomes moresevere.

[0040] The present invention effectively counteracts this by means ofthe measures described above by providing a means for changing thefrequency response of the motor with respect to ground potential suchthat there are now virtually no pronounced resonant peaks f_(res) in thevicinity of the natural system frequencies f_(sys) of the convertersystem shown in FIG. 1.

BRIEF DESCRIPTION OF THE DRAWINGS

[0041] Further details and advantages of the invention will be apparentfrom the following description of an exemplary embodiment and inconjunction with the drawings, wherein elements having the samefunctionality are denoted by the same reference symbols. In the figures:

[0042]FIG. 1 shows a block diagram of a converter system having athree-phase motor using a converter with an intermediate voltage circuitand a controlled input converter, and a supply network input inductor inthe step-up converter mode;

[0043]FIG. 2 shows an equivalent circuit of the passive circuit, formedby the arrangement of a converter system shown in FIG. 1, with regard tosystem oscillations;

[0044]FIG. 3 shows an outline sketch of a lattice network structureformed in a motor;

[0045]FIG. 4 shows a block diagram system model of the effective path ofthe voltages with respect to ground potential from the supply network tothe motor star point;

[0046]FIG. 5 shows a schematic block diagram of a topology in aconverter system;

[0047]FIG. 6 shows an outline sketch of a balanced drive for the motorcomprising the intermediate voltage circuit on the basis of two phasesL1 and L2;

[0048]FIG. 7 shows a timing diagram of the voltage profile between thesetwo phases L1 and L2, compared to the switching states of the inverterW;

[0049]FIG. 8 shows a timing diagram of the voltage profile of the phaseL1 with respect to ground;

[0050]FIG. 9 shows a corresponding timing diagram of the voltage profileof the phase L2 with respect to ground;

[0051]FIG. 10 shows an outline sketch of an unbalanced drive for themotor comprising the intermediate voltage circuit as a common-modesystem for analysis of phase to ground;

[0052]FIG. 11 shows a timing diagram of the unbalanced voltage profileof the phases L1 and L2 with respect to ground;

[0053]FIG. 12 shows a corresponding timing diagram of the DC componentof the unbalanced voltage profile of the phases L1 and L2 with respectto ground;

[0054]FIG. 13 shows a corresponding timing diagram of the AC componentof the unbalanced voltage profile of the phases L1 and L2 with respectto ground;

[0055]FIG. 14 shows an amplitude/frequency response of a motor withrespect to ground in order to illustrate the transfer function H₂(s);

[0056]FIG. 15 shows an amplitude/frequency response of a motor withrespect to ground ignoring the natural damping, which increases as thefrequency rises, in order to illustrate the transfer function H₂(s);

[0057]FIG. 16 shows an outline sketch of a lattice network structureformed from four-pole networks;

[0058]FIG. 17 shows an example of a winding layout of a motor windingusing field coil technology;

[0059]FIG. 18 shows a cross-sectional view of the installed position ofthese field coils in the laminated core;

[0060]FIG. 19 shows the unbalanced equivalent circuit of such anarrangement as shown in FIG. 17 and FIG. 18;

[0061]FIG. 20 shows the same lattice network structure as that in FIG. 3with the transformer-coupled damping resistance according to theinvention;

[0062]FIG. 21 shows an equivalent circuit for the coupling core and theimpedance as shown in FIG. 20; and

[0063]FIG. 22 shows a comparison of the amplitude/frequency response(amplitude profile plotted against the frequency) with and without thetransformer-coupled damping resistance.

DETAILED DESCRIPTION OF THE INVENTION

[0064]FIG. 1 to FIG. 3 have been explained in the introduction in orderto assist in understanding the problems which the present inventionaddresses, although it should again be mentioned that the recognition ofthe problem of system oscillations in a converter system as shown inFIG. 1, particularly with a supply network-side input inductor L_(K) inthe step-up converter mode in conjunction with a motor with a latticenetwork structure K, and the cause of that problem is not known from theprior art. Hence, the recognition of the problem represents an importantaspect of the present invention.

[0065] The system model of a converter system as shown in FIG. 1 will beanalyzed with regard to an effective path from the supply network to themotor star point. To this end, FIG. 4 shows the input-side supplyvoltage U_(N) with respect to ground, converted via the convertersystem, which has a first transfer function H₁(s), to the voltageU_(P600) on the positive intermediate circuit rail with respect toground. In the motor, voltage U_(P600) is converted via a secondtransfer function H₂(s) to a voltage U_(S) with respect to ground whichis present at the motor star point S.

[0066] In practice, a number of motors are frequently operated using oneconverter system, by a number of inverters W₁ to W₃ with motors M₁ to M₃connected to them being supplied from the intermediate circuit voltageU_(ZK), as shown in FIG. 5. The input converter E is supplied from thesupply network N via the filter arrangement F and supplies a number ofinverters W₁ to W₃, with motors M₁ to M₃ connected to them, from theintermediate circuit voltage U_(ZK).

[0067] Between the respective inverters W₁ to W₃ with connected motorsM₁ to M₃, it must be remembered, with regard to system oscillations,that there is a natural system frequency f_(sys), which describes theresonant frequency f_(res)(sys) of the system, with N, F, E, W₁ to W₃ onthe converter system side. In contrast, the motors M₁ to M₃ have theirown resonant frequency f_(res), which corresponds to the naturalfrequency f_(res)(mot) of the respective motor.

[0068] The theoretical system analysis shown in FIG. 4 is thus carriedout separately for each respective motor, for which reason the transferfunction H₁(s) represents the effect of the filter F, the inductanceL_(K), the input converter E, all the inverters W, all the other motorsM and all the lines LT. Consequently, all of the elements in the dashedbox shown in FIG. 5 are modeled as H₁(s) when analyzing the behavior ofmotor M3.

[0069] A system oscillation which is excited in particular by thepulsing of a supply E and, to a lesser extent, also by the pulsing ofthe inverters W in the shaft modules, can be formed in such a converteror converter system. This pulsing results in periodic changing of thecharge on the parasitic capacitances, as has been described withreference to FIG. 1.

[0070] If the supply network voltage U_(N) is regarded as an inputvariable, then this is mapped by the transfer function H₁(s) onto theoutput variable U_(P600) (when considering the positive intermediatecircuit rail P600). Except for 600 V DC components, the voltage U_(P600)is applied in the common mode to the motor terminals, and thiscorresponds to an unbalanced system or zero phase-sequence system.

[0071] The motor line LT can in theory be associated with both H₁(s) andH₂(s). In this case, as mentioned, the motor line LT will be associatedwith H₁(s). In the frequency band under consideration, the line LT canbe regarded as an electrical short.

[0072] As already mentioned, the passive circuit formed in this way andillustrated in FIG. 2 has a natural resonant frequency f_(res)(sYs) orf_(sys), at which this system starts to oscillate. Accordingly, inaddition to the voltage variations with an amplitude of 600 V, forexample, that occur during operation, an additional, undesirableoscillation with an amplitude of up to several hundred volts is alsomodulated onto the voltages on the intermediate circuit rails P600 andM600. This means that the output voltages from the inverter W withrespect to ground are no longer in step-function form, as is the casebetween two phases U, V, W, with the output voltages now including thesystem oscillations present on the intermediate circuit rails P600 andM600.

[0073] This is illustrated in FIG. 6, which shows a balanced drive forthe motor M from the intermediate voltage circuit C_(ZK) on the basis oftwo phases L1 and L2, by way of example. FIG. 6 shows the intermediatecircuit with the intermediate circuit capacitance C_(ZK) and theintermediate circuit rails P600 and M600, from which, using a simplifiedinverter in the form of a bridge circuit with the switches S1 to S4, avoltage U_(L1L2) or a current i is produced in order to supply twosections L1 and L2 of the motor M, which are connected at the motor starpoint S, each having inductances L_(H). The motor has the previouslydescribed discharge capacitance C_(M) with respect to ground potential.

[0074]FIG. 7 shows the profile of the voltage U_(L1L2) between thephases L1 and L2 plotted against time t and compared with the respectiveswitching states of the switches S1 to S4 in the bridge of the inverterW, which are plotted underneath, likewise with respect to time t. Theswitches S1 and S2 represent the first bridge arm, and the switches S3and S4 represent the second bridge arm. In this case, switches in onephase are always in opposite directions to one another since, otherwise,the intermediate circuit would be short-circuited.

[0075] The four states 1, 2, 3 and 4 are assumed in order to illustratethe switching states of the two bridge arms, S1/S2 and S3/S4. In state1, S1=0, S2=1 and S3=0, S4=1 with the state “−−” for the phases L1 andL2. Thus, in this case, what are referred to as zero vectors NZ areswitched and there is no voltage U_(L1L2) between the phases L1 and L2.In state 2, S1=1, S2=0 and S3=0, S4=1. This results in the state “+−”,with a voltage U_(L1L2) of 600 V between the phases L1 and L2. In state3, S1=1, S2=0 and S3=1, S4=0. This results in the state “++”, that is tosay zero vectors NZ are once again switched, and there is no voltageU_(L1L2) between the phases L1 and L2. Finally, in state 4, S1=0, S2=1and S3=1, S4=0. This results in the state “−+” with a voltage U_(L1L2)of −600 V between the phases L1 and L2. A new state 1 then starts, etc.

[0076]FIG. 8 also shows the voltage of phase L1 with respect to ground,plotted against time t for these states 1 to 4, that is to sayconsidered asymmetrically. In this case, the phenomenon described abovecan be seen, as a result of which the voltage profile does not have anideal step-function form since it is modulated with the undesirablesystem oscillations of the converter system from FIG. 1 and FIG. 4 withan amplitude, for example, of approximately 150 V. The same applies, insome circumstances, to a constant amplitude shift for the unbalancedvoltage of phase L2 with respect to ground, which is shown in FIG. 9. Ascan be seen, both phases L1 and L2, and thus also the intermediatecircuit rails P600 and M600, oscillate in time with one another. Thismeans that they are always shifted “in parallel”, that is to say withoutphase shift.

[0077] It is clear from this that the problem of resonant peaks iscaused by unbalanced currents i. For this reason, it is worthwhileanalyzing the arrangement as a common-mode system, a section of which isshown in FIG. 10 in the form of an unbalanced drive for the motor M fromthe intermediate voltage circuit C_(ZK). In this case it is assumed thatall the motor phases U, V, W or L1 to L3 form an inductance L_(σ) whichis governed by the motor winding and is terminated by the dischargecapacitance C_(M) to ground.

[0078] If one now considers the two phases L1 and L2 jointly, as acommon-mode system (referred to as L1&&L2 in the following text) thenthis results in the voltage with respect to ground over time shown inFIG. 11. No common-mode signal can be shown in the states 2 and 4 in thecommon-mode system for L1&&L2 from the “parallel” shift of theindividual phases L1, L2 which can be seen in FIG. 8 and FIG. 9, sincein this case the phases L1 and L2 are at a different potential (in thesketched situation, the DC voltage difference is 600 volts). Since onlytwo phases are considered, they produce an average of zero volts in thecommon mode. In the other states 1 and 3, the voltage of L1&&L2 overtime corresponds to that of L1 in FIG. 8, and L2 in FIG. 9.

[0079] The voltage of L1 &&L2 over time as shown in Figure II in thecommon-mode system can in this case be separated into a fundamental GWand a harmonic OW. These are shown separately in FIG. 12 and FIG. 13.The illustration in FIG. 12 shows the voltage of the fundamental GW overtime. This clearly shows that this voltage profile describes the desiredstep-function form switching states with −300 V in state 1, 0 V instates 2 and 4 owing to the “parallel” shift, and +300 V in state 3. Theharmonic OW of the voltage L1&&L2, shown over time in FIG. 13, describesan essentially constant sinusoidal profile with an amplitude of, forexample, 150 V.

[0080] The harmonic or system oscillation is thus applied to the motor Min all states 1 to 4, as a result of which this phase-ground tunedcircuit is always excited in the motor M, as shown in FIG. 2. If thissystem oscillation is now in the vicinity of a natural motor frequency,and/or the motor M has a pronounced resonance in the vicinity of thefrequency of the system oscillation, undesirable resonant peaks canoccur. A “maximum” oscillation amplitude of this phase-ground tunedcircuit is generally prevented only by the discontinuities in theharmonic resulting from the switching from one state to the next.

[0081] With regard to the theoretical system analysis of the problemsshown in FIG. 4, as mentioned above, the amplitude of such a systemoscillation f_(sys) in this case depends essentially on two factors.Namely, the intrinsic damping in the system, which is inverselyproportional to the Q-factor of the tuned circuit with the dampingincreasing as the frequency rises, and the excitation, that is to saythe nature of the supply (for example diode supply or regulated supply)and the magnitude of the intermediate circuit voltage U_(ZK).Particularly pronounced natural system oscillations can thus be observedin converter systems which have a large number of shaft modules W andmotors M, and long motor lines LT. The frequency band of the naturalsystem oscillations f_(sys) in this case generally extends fromapproximately 10 kHz for large converter systems up to more than 50 kHzfor smaller converter systems.

[0082] The amplitude and frequency thus depend on the configuration andthe extent of the system, for example:

[0083] the nature of the supply E (regulated or unregulated);

[0084] the number of shafts or motors M which are operated using oneconverter system UR; and

[0085] the length of the power lines LT.

[0086] It should thus be stated at this point that converter systemswith an intermediate voltage circuit may have natural oscillations onthe intermediate circuit rails P600, M600 with respect to ground. Theseare particularly pronounced in multi-shaft systems and with a regulatedinput converter E, particularly when used in the step-up converter mode.The motor M in an unbalanced system is thus excited virtually at asingle frequency irrespective of the pulse patterns on the individualphases U, V, W or L1 to L3. The transfer function H₂(s) results in thisexcitation being mapped onto the output side, namely the voltage Us atthe star point S with respect to ground.

[0087] All electric motors M, irrespective of the type, qualitativelyhave a transfer function H₂(s) with respect to ground whoseamplitude/frequency response A(f) is as shown in the illustration inFIG. 14. This has a pronounced resonant frequency f_(res)(mot) orf_(res). The transfer function H₂(s) can in this case be described as:

H ₂(s)=U _(P600) /U _(S)

[0088] The frequency of the pronounced resonant peak of the motordepends on the inductive and capacitive elements in the motor withrespect to ground, and is thus governed by:$f_{res} \propto \frac{1}{\sqrt{L_{M} \cdot C_{M}}}$

[0089] where L_(M)=f(L_(PE)) is the effective inductance andC_(M)=f(C_(PE)) is the effective capacitance of the motor M with respectto ground potential PE or in the zero phase-sequence system. The precisefunctions in this case depend on the respective measurement method andon the equivalent circuits used.

[0090] If there are a number of star points S, identical tuned circuitsare connected in parallel. The capacitance per tuned circuit is thengoverned by: $\overset{\sim}{C} \propto {\frac{C_{M}}{{number}_{S}}.}$

[0091] The inductance depends on the number of series-connected coils,with there being a number of star points S, particularly when usingfield coil technology. Since the individual coils can be regarded asbeing magnetically decoupled from one another, it can furthermore bestated that: $\overset{\sim}{L} \propto \frac{n_{1S}}{{number}_{S}}$

[0092] where n_(1s) is the number of coil assemblies for one star pointS, and number_(s) is the number of star points S.

[0093] Thus, for motors of the same physical size but in which identicalcoil groups are connected differently:$f_{res} \propto \frac{1}{\sqrt{\frac{1}{{number}_{S}} \cdot \frac{1}{{number}_{S}}}} \propto {{number}_{S}.}$

[0094] The influence of the motor size on the resonant frequency f_(res)can be estimated as follows: $C = \frac{ɛ \cdot A}{d}$

[0095] where

A∝slot area∝D·LG

[0096] where D is the diameter and LG the length of the motor.

[0097] The influence of the motor size with the characteristicsotherwise being constant is thus reflected by:$f_{res} \propto \frac{1}{\sqrt{{slot}\quad {area}}} \propto {\frac{1}{\sqrt{D \cdot {LG}}}.}$

[0098] Ignoring the natural damping, which increases as the frequency frises, resulting from eddy current losses, hysteresis etc. and inparticular if the motor M is regarded as a lattice network K, as appearsto be macroscopically plausible particularly in the case of motors usingfield coil technology since the coil groups are connected in series,this results in the amplitude/frequency response A(f) shown in FIG. 15.The response graph A(f) has a number of local maxima which describe anumber of resonant frequencies f_(res 1) to f_(res n), with the firstresonant peak f_(res 1), which occurs at the lowest frequency, beingdominant and thus representing the governing or pronounced resonantfrequency f_(res).

[0099] There are thus frequencies, particularly the lowest resonantfrequency f_(res), at which considerably greater voltages occur at themotor star point S than at the input terminals of the motor M and theseare, for example, greater by a factor of 3 to 4. In this case, it can bestated that the resonant peak becomes higher the lower f_(res) is.Geometrically large torque motors are thus particularly at risk, inwhich resonant points f_(res) which are in the vicinity of or areprecisely at the frequency f_(sys) of the natural system oscillationscan be formed particularly easily due to the slot area and the number ofstar points S.

[0100] This knowledge is important in considering the lattice networkstructure K as shown in FIG. 3. This is because such a lattice networkstructure K can be regarded not only macroscopically in motors usingfield coil technology, but also, in principle, with other types. To thisend, the motor M together with its motor winding can also be regarded inentirely general form as a microscopic lattice network K composed ofidentical four-pole networks V1 to Vn, as is shown in the illustrationin FIG. 16. Each four-pole network V1 to Vn in this case comprises aninductance L, which is connected in series with a non-reactiveresistance R. The output voltage is in this case dropped in parallelwith a capacitance C, which is connected as a voltage divider with L andR.

[0101] In order to illustrate this structure, the lattice networkstructure K will be described in more detail. To this end, FIGS. 17 to19 show a comparison of the construction of a motor winding using fieldcoil technology on the basis of its unbalanced electrical equivalentcircuit. The illustration in FIG. 17 shows a coil group in the phase U.Each coil group of the motor winding MW, comprising a number ofseries-connected field coils PS1 to PS3, forms with respect to ground amacroscopic LC lattice network with an inductance L and a capacitance C.The start and end of this lattice network K are the input terminal U andstar point S of the motor M. As described above, this lattice network Khas a number of resonant frequencies f_(res).

[0102] If such a lattice network is excited at the input U (phase toground) for example sinusoidally at its lowest natural frequencyf_(res), which is generally the most pronounced, then it can supply aconsiderably higher voltage at the star point S than at the input U. Inthe worst case, this voltage can lead to a breakdown in the maininsulation in the vicinity of the star point S. Such sinusoidalexcitation can occur in particular as a result of inadvertent systemoscillation f_(sys) (as described above) of the entire converter system.

[0103] The described mechanism is most clearly pronounced when at restsince, in this case, all the phases U, V, W or L1, L2, L3 are switchedat the same time. The natural frequencies and damping levels in thelattice network K depend on the construction of the motor winding, forexample, on:

[0104] the number of coils per path

[0105] the number of turns (inductance)

[0106] the shape of the coils (capacitance)

[0107] the encapsulation (capacitance).

[0108] The illustration in FIG. 18 shows how the unbalanced equivalentcircuit of such an arrangement as shown in FIG. 19 is obtained from anexample of a winding design as shown in FIG. 17 and the installedposition of the field coils PS1 to PS3 in the laminated core B.

[0109] The winding organization means that the first layer (solidcircles) of a field coil PS1 to PS3 always has a larger capacitance Cwith respect to the laminated core B (ground potential) than the otherlayers (hollow circles).

[0110] The respective inductance L is formed by the respective fieldcoil PSI, PS2, PS3 itself, on the assumption that the mutual inductancebetween the coils can be ignored since, at the frequencies underconsideration (for example 20 kHz), the iron only slightly amplifies andguides the magnetic flux. This is confirmed by the fact that theomission of the secondary part scarcely changes the measured values(frequency, amplitude).

[0111] The illustrated structure thus has as large a number of latticenetwork elements (n=3) as field coils PS1 to PS3. If a winding sectioncomprises m parallel-connected paths (lattice structures), then m=3paths are connected in parallel for operation with a zero vector NZ(simultaneous switching of all the input terminals) for the phases U, V,W.

[0112] The natural damping in the system is not shown in FIG. 19.Initially, this should be regarded as the non-reactive resistance R ofeach inductance L.

[0113] In the following example, the transfer function of the latticenetwork structure which is shown in FIG. 16 and comprises the four-polenetworks V1 to Vn from the phase terminals to the star point S iscalculated for a motor M by using the variables L_(Z), C_(M), m and n.

[0114] In this case, C_(M) describes the winding-ground capacitance, andL_(Z), R_(Z) describe the impedance of one winding path. The expressionwinding path in this case means the series-connected coils from onephase terminal U, V or W to a star point S, with any parallelconnections which exist within a winding section being disconnected.

[0115] This allows the parameters for a lattice network element orfour-pole network to be determined as follows: $L = \frac{L_{Z}}{n}$$C = \frac{C_{M}}{3 \cdot n \cdot m}$

[0116] m: number of parallel paths;

[0117] n: number of coils in series.

[0118] The loss resistance R of the inductance L is initially set to$R = \frac{R_{Z}}{n}$

[0119] with the value for R_(Z) in the series model being predetermined,for example, 10 kHz.

[0120] This value can be only a first approximation, since R is highlyfrequency-dependent and is determined at a lower frequency than thefrequencies which actually occur.

[0121] The Z-matrix of an individual four-pole element Z_(v) is:${\underset{\_}{Z}}_{v} = \begin{bmatrix}{R + {sL} + \frac{1}{sC}} & \frac{1}{sC} \\\frac{1}{sC} & \frac{1}{sC}\end{bmatrix}$

[0122] The lattice matrix A_(v) can be formed as follows:${\underset{\_}{A}}_{v} = \begin{bmatrix}{{s^{2}{LC}} + {sRC} + 1} & {{sL} + R} \\{sC} & 1\end{bmatrix}$

[0123] The A-matrix for the entire lattice network K is then:

A _(tot) =A _(v) ^(n)

[0124] with the element${{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack} = \frac{{\underset{\_}{U}}_{1}}{{\underset{\_}{U}}_{2}}$

[0125] The amplitude/frequency response A(s) is thus, as a complexgeneralization of A(f):${A(s)} = {20\quad 1\quad {g\left( {\frac{1}{{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack}} \right)}\quad {in}\quad {db}}$

[0126] If it is assumed that the damping resistance R increases withfrequency, then the higher resonant frequencies would be even morestrongly damped.

[0127] If the A-matrix for the entire lattice network K is used to formthe Z-matrix, then the element Z_(in) is equal to the input impedance ofthe lattice network.

[0128] This results in the input impedance, as follows:$Z_{in} = {{\frac{{\underset{\_}{Z}}_{tot}\left\lbrack {1,1} \right\rbrack}{m \cdot n}} = {\frac{1}{m \cdot n} \cdot {\frac{{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack}{{\underset{\_}{A}}_{tot}\left\lbrack {2,1} \right\rbrack}}}}$

[0129] The illustration in FIG. 20 shows a lattice network structure Kformed in a motor M, as has been explained with reference to FIG. 3, butnow with an impedance Z according to the invention which istransformer-coupled into all three phases U, V, W of the motor M. Thisis done, for example, by routing each phase U, V, W of the motor M,whose winding section represents a lattice network K, through one andthe same magnetic core MK, which is either sufficiently lossy owing toits material characteristics or else has a winding which is electricallyshort-circuited by an impedance Z (as in FIG. 20). Since all the phasesare routed through one magnetic core MK, only the unwanted common-modeprocesses are damped with respect to the natural system oscillations inthe converter system.

[0130]FIG. 21 shows the transformer equivalent circuit relating to thearrangement shown in FIG. 20. The magnetic core MK is in this casedescribed by the four-pole network equivalent circuit of a transformerwith the main inductance of the transformer and the total inductance ofthe coupling coil L_(H1), together with the stray inductances L_(σ1) andL_(σ2). The impedance Z is in this case composed of a series circuitformed by a non-reactive damping resistance R_(a) and a parasiticinductance L_(R), which is in general formed by the supply lines of theresistance. The impedance Z may, of course, also include other elementssuch as capacitances, which are used to compensate for L_(σ2) and L_(R).

[0131] On the basis of this equivalent circuit for the coupling core MKand the impedance Z, R_(a) and L_(H1) can now be dimensioned so that asufficiently large time constant is achieved resulting in a satisfactorydamping effect with minimized losses in the impedance Z. To ensure thisresult, the time constant should correspond to approximately half thelowest motor resonant frequency f_(res).

[0132] The resistive component R_(a) is preferably dimensioned inaccordance with$R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}$

[0133] In this case, L describes the inductance and C the dischargecapacitance of a four-pole network V1 . . . Vn, or of a lattice networkLC element as illustrated in FIG. 16 or FIG. 19.

[0134] The main inductance of the transformer and the total inductanceof the coupling core MK are preferably dimensioned in accordance with

[0135] where${L_{Hl} \geq \frac{R}{2\pi \quad f_{0}}},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

[0136] and f_(res) is the pronounced resonant frequency of the motor M.

[0137] It can be seen that the inductance L_(H1) should not be reducedby saturation effects. For this reason, the coupling core MK should bedimensioned such that it cannot enter saturation at any operatingpoints.

[0138]FIG. 22 shows the amplitude/frequency response A(f) with respectto ground potential with and without the impedance Z beingtransformer-coupled according to the invention into all three phases U,V, W of the motor M. The solid line corresponds to the undampedsituation as shown in FIG. 15, with the first resonant frequency as themost pronounced resonant frequency f_(res). Transformer damping of theunwanted common-mode processes, dimensioned as described above, resultsin the dotted profile, where the resonant peak at f_(res) isconsiderably lower. Thus, if this is located in the vicinity of thenatural system frequency f_(sys) of the converter system, then there isno need to be concerned about resonant peaks with the described negativeconsequences.

[0139] The advantages of the invention are a reduction in the load onthe insulation system with respect to ground, which improves thereliability and robustness of the motor and elimination of the need forexpensive additional motor insulation that reduces the rating of themotor, since the voltage load remains in areas which are regarded ascapable of handling the voltage load using standard materials based onthe present prior art for low-voltage motors.

[0140] The foregoing merely illustrates the principles of the inventionin exemplary embodiments. Various modifications and alterations to thedescribed embodiments will be apparent to those skilled in the art inview of the teachings herein. For example, the principles of theinvention could be applied if there are a number of motor star points S,as is the situation, for example, in linear motors or torque motors. Tothis end, a suitable impedance Z for each motor star point S is in eachcase transformer-coupled into all the motor phases U, V, W leading to anindividual motor star point S. In principle, resonant peaks also existin motors using what is referred to as wild winding (standard forlow-voltage motors), so that the principles of the invention could beused for these motors and for motors other than the field coiltechnology motors chosen for illustrative purposes. It will thus befully appreciated that those skilled in the art will be able to devisenumerous systems and methods which, although not explicitly shown ordescribed, embody the principles of the invention and thus are withinthe spirit and scope of the invention as defined in the appended claims.

We claim:
 1. A method for damping resonant electrical peaks at a motorstar point in an electric motor which is operated using an intermediatevoltage circuit converter with an input-side inductance, which, owing tocharacteristics of its winding sections, has a frequency response withat least one resonant frequency with respect to ground potential,comprising: introducing into all phase paths leading to at least onemotor star point, an impedance for damping capacitive discharge currentsto ground potential, which are caused by system oscillations excitedasymmetrically with respect to ground in the motor phases of theconverter system in the winding sections.
 2. The method according toclaim 1, wherein all the motor phase paths leading to the at least onemotor star point are routed through a lossy magnetic coupling core. 3.The method according to claim 2, wherein the coupling core has a windingthat is short-circuited via an impedance.
 4. The method according toclaim 2, wherein said lossy coupling core is coupled to the input to themotor.
 5. The method according to claim 1, wherein the impedancecomprises a non-reactive resistance.
 6. The method according to claim 5,wherein each winding section of the motor forms an LC lattice network,and the non-reactive resistance is dimensioned in accordance with theformula:${R_{d} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}},$

where L is the inductance and C the discharge capacitance of one latticenetwork element in the LC lattice network structure.
 7. The methodaccording to claim 5, wherein the total inductance of the couplingcircuit formed with the magnetic core is dimensioned in accordance withthe formula: where, preferably,${L_{H1} \geq \frac{R}{2\pi \quad f_{0}}},{{where}\quad {preferably}},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

with f_(res) being a pronounced resonant frequency.
 8. The methodaccording to claim 1, wherein the converter is operated together withthe input-side inductance in order to provide a supply based on thestep-up converter principle.
 9. The method according to claim 1, whereinthe input-side inductance comprises an inductor.
 10. An electric motorfor operation using an intermediate voltage circuit converter having aninput-side inductance and having a frequency response that is governedby winding inductances and discharge capacitances, with a pronouncedresonance with respect to ground potential, in which all the motor phaselines leading to a motor star point are routed through a lossy magneticcoupling core.
 11. The electric motor according to claim 10, wherein thecoupling core has a winding which is short-circuited via an impedance.12. The electric motor according to claim 10, wherein the coupling coreis coupled to at the input to the motor.
 13. The electric motoraccording to claim 11, wherein the impedance is so pronounced that itdamps capacitive discharge currents to ground potential, which arecaused by system oscillations excited asymmetrically with respect toground in the motor phases of the converter system in the windingsections.
 14. The electric motor according to claim 11, wherein theimpedance comprises a non-reactive resistance.
 15. The electric motoraccording to claim 14, wherein each winding section of the motor formsan LC lattice network, and the non-reactive resistance is dimensioned inaccordance with the formula:${R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}},$

where L is the inductance and C the discharge capacitance of one latticenetwork element in the LC lattice network structure.
 16. The electricmotor according to claim 14, wherein the total inductance of thecoupling circuit formed with the magnetic core is dimensioned inaccordance with the formula: where, preferably,${L_{H1} \geq \frac{R}{2\pi \quad f_{0}}},{{where}\quad {preferably}},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

with f_(res) being a pronounced resonant frequency.
 17. The electricmotor according to claim 10, wherein the coupling core is dimensioned insuch a manner that it does not enter saturation at any operating pointof the motor.
 18. The electric motor according to claim 10, furthercomprising winding sections using field coil technology, wherein eachwinding section form a lattice network structure composed of inductancesand discharge capacitances, in which case the impedance is used fortransformer damping of these lattice network structures.
 19. Theelectric motor according to claim 15, wherein the impedance is sopronounced that it damps common-mode currents excited asymmetricallywith respect to ground in the motor phases of the converter system inthe lattice network structure.
 20. An electrical drive according toclaim 10, wherein said motor employs wild winding technology, inparticular a low-voltage motor, which has low resonant frequencies byvirtue of its construction.